Various interference influences which must be taken into account in the signal detection at the receiver end occur during the transmission of radio signals between a transmitter and a receiver. Firstly, the signal is subject to distortion which is caused by there generally being two or more possible signal propagation paths. Reflection, scatter and diffraction of signal waves on obstructions such as buildings, mountains and the like result in the received field strength at the receiver being composed of a number of signal components which are generally subject to different delay and have different intensities. This phenomenon, which is referred to as multipath propagation, causes the distortion of the transmitted data signal which is known as intersymbol interference (ISI).
Other active subscribers represent a further cause of interference. The interference caused by these subscribers is referred to as multiple access interference (Multi Access Interference: MAI). A situation frequently occurs in which a dominant interference source or interference channel actually has a serious adverse effect on the signal detection in the payload channel.
First of all, only one channel will be considered, that is to say MAI will be ignored. This multipath transmission channel between the transmitter S and the receiver E can be modelled as a transmission filter H with a channel coefficient hk as is illustrated in FIG. 1. The transmitter S feeds transmission symbols sk into the transmission channel, that is to say the channel model transmission filter H. An additive noise contribution nk, which is added to the transmission symbols sk which have been filtered with hk at the output of the channel model transmission filter H, can be taken into account by a model adder SU.
The index k denotes the discrete time in time units of the symbol clock rate. The transmission signals sk, which have been filtered by the transmission filter H and on which noise has been superimposed are received as the received signal xk by the receiver E, in which case:
                              x          k                =                                            ∑                              i                =                0                            L                        ⁢                                                  ⁢                                          h                i                            ⁢                              s                                  k                  -                  i                                                              +                      n            k                                              (        1        )            where L represents the order of the transmission channel being modelled by the filter H. As can be seen from equation (1), ISI is present since xk depends not only on sk but also on sk−1, . . . ,sk−L.
FIG. 2 shows the channel model transmission filter H. The filter H comprises a shift register composed of L memory cells Z. There are taps (a total of L+1 of them) before and after each memory cell Z, which lead to multipliers which multiply the values of the symbols sk,sk−1 . . . , sk−L (which are inserted into the shift register via an input IN at the symbol clock rate T−1 by the corresponding channel impulse responses h0,h1, . . . ,hL. An output stage AD of the filter H adds the outputs of the L+1 multipliers, thus resulting in an output signal OUT in accordance with equation 1.
The memory contents of the channel model shift register describe the state of the channel. The memory contents of the first memory cell on the input side contain the symbol sk−1 (which is multiplied by h1) in the time unit k, and the other memory cells Z are filled with the symbols sk−2,sk−3, . . . ,sk−L. The state of the channel in the time unit k is thus determined unambiguously by the details of the memory contents, that is to say by the L−tuple (sk−L,sk−L+1, . . . ,sk−1).
The received signal values xk in the receiver E are known as sample values, and the channel impulse responses h0,h1, . . . ,hL of the channel are estimated at regular time intervals. The equalization task comprises the calculation of the transmission symbols sk from this information. The following text is based on equalization by means of a Viterbi equalizer.
Viterbi equalization is based on finding the shortest route through a state diagram of the channel, which is known as the trellis diagram. The channel states are plotted against the discrete time k in the trellis diagram. According to the Viterbi algorithm (VA), a branch metric which represents a measure of the probability of a transition is calculated for each possible transition between two states (predecessor state relating to the time unit k→destination state relating to the time unit k+1). The branch metrics are then added to the respective state metrics (which are frequently also referred to in the literature as path metrics) of the predecessor states. In the case of transitions in the same final state, the sums obtained in this way are compared. That transition to the final state in question whose sum of the branch metric and state metric of the predecessor state is minimum is selected and forms the extension of the path leading from this predecessor state to the destination state. These three basic VA operations are known as ACS (ADD-COMPARE-SELECT) operations.
While, from the combinational point of view, the number of paths through the trellis diagram increases exponentially as k increases (that is to say as time passes), it remains constant in the case of the VA. The reason for this is the selection step (SELECT). Only the selected path (survivor) survives and can be continued. The other possible paths are rejected. Recursive path rejection is the core concept of VA and is an essential precondition of computationally coping with the problem of searching for the shortest path through the trellis diagram.
The number of channel states (that is to say the number of ways in which the shift register H may be filled) in the trellis diagram, which is identical to the number of paths followed through the trellis diagram, is pL. In this case, p denotes the significance of the data symbols being considered. The computation complexity of VA accordingly increases exponentially with L. Since L should correspond to the length of the channel memory of the physical propagation channel, the complexity for processing the trellis diagram increases as the channel memory of the physical propagation channel increases.
One simple method for reducing the computation complexity is to base the trellis processing on a short channel memory L. However, this has a severe adverse influence on the performance of the equalizer. A considerably more sensible measure for limiting the computation complexity, and which does not have such a serious influence on the quality of the equalizer, is the decision feedback (DF) method. In the DF method, the VA is based on a reduced trellis diagram, that is to say a trellis diagram in which only some of the pL channel states are considered, rather than all of them. If the trellis diagram is reduced to pLDF trellis states (LDF<L), the remaining L−LDF channel coefficients (which are not used for the definition of trellis states) are still taken into account by using them for calculation of the branch metrics in the reduced trellis diagram.
A branch metric between two states must be calculated for each possible transition both during the processing of the complete trellis diagram and of the processing of the reduced trellis diagram (DF situation). The branch metric is the Euclidean distance between the measured signal value or sample value Xk and an estimated “hypothetical” signal value, which is calculated and “tested” with respect to the destination state, the transition from the predecessor state to the destination state, and the path history, taking into account the channel knowledge in the receiver.
In order to explain this, let us assume by way of example that p=2 (binary data signal), that is to say there are 2L (DF situation: 2LDF) trellis states (0,0, . . . ,0), (1,0, . . . ,0) to (1,1 . . . , 1) comprising L tuples (DF: LDF−tuples). A specific hypothetical predecessor state is assumed to be defined by the shift register occupancy (aL,aL−1, . . . ,a1) (in the DF situation, only the LDF right-hand bits (aLDF, . . . , a1) of the shift register occupancy are used for the state definition). The hypothetically transmitted symbol (bit) 0 or 1 which leads from the predecessor state (aL, aL−1, . . . , a1) in the time step k to the destination state (aL−1,aL−2, . . . ,a0) in the time step k+1 (DF: predecessor state (aLDF, . . . , a1) to the destination state (aLDF−1, . . . ,a0) is annotated a0. With or without DF, the branch metric BMk is:BMk=|sample value−estimated signal value|2 
                    =                              |                                          X                k                            -                              (                                                                            ∑                                              i                        =                        1                                            L                                        ⁢                                                                                  ⁢                                                                  h                        i                                            ⁡                                              (                                                  1                          -                                                      2                            ·                                                          a                              i                                                                                                      )                                                                              +                                                            h                      0                                        ⁡                                          (                                              1                        -                                                  2                          ·                                                      a                            0                                                                                              )                                                                      )                                      ⁢                          |              2                        ⁢                          for              ⁢                                                          ⁢                              a                i                                              =                      {                          0              ,              1                        }                                              (        2        )            
The estimated signal value (also referred to in the following text as the estimated symbol) is the sum of products of a channel coefficient and a symbol. For the DF situation, the term
      ∑          i      =      1        L    ⁢          ⁢            h      i        ⁡          (              1        -                  2          ·                      a            i                              )      may also be split into a trellis contribution and a DF contribution:
                              BM          k                =                  |                                    X              k                        -                          (                                                                                          ∑                                              i                        =                                                                              L                            DF                                                    +                          1                                                                    L                                        ⁢                                                                                  ⁢                                                                  h                        i                                            ⁡                                              (                                                  1                          -                                                      2                            ·                                                          a                              i                                                                                                      )                                                                                                  ︸                                          DF Contribution                                                                      +                                                                            ∑                                              i                        =                        1                                                                    L                        DF                                                              ⁢                                                                                  ⁢                                                                  h                        i                                            ⁡                                              (                                                  1                          -                                                      2                            ·                                                          a                              i                                                                                                      )                                                                                                  ︸                                          Trellis contribution                                                                      +                                                                            h                      0                                        ⁡                                          (                                              1                        -                                                  2                          ·                                                      a                            0                                                                                              )                                                                            ︸                                          hyp. Symbol contribution                                                                                  )                                ⁢                      |            2                                              (        3        )            
This means that the estimated symbol comprises two (DF situation: three) contributions: a contribution which is determined by the hypothetically transmitted symbol a0 for the transition from the time unit k to the time unit k+1, the trellis contribution, which is given by the predecessor state relating to the time unit k in the trellis diagram, and, in the DF situation, the reduced trellis states also result in the DF contribution.
The branch metric BMk is always the same, with or without DF. The computation saving in the case of VA with DF results, as already mentioned, from the smaller number 2LDF of trellis states to be taken into account in the processing of the trellis diagram, that is to say from the reduction in the trellis diagram.
Furthermore, if it is intended to take into account an interference channel (that is to say a second multipath transmission channel) in the equalization of a data signal, joint VA equalization must be carried out on both channels (the payload channel and the interference channel). An overall trellis diagram is constructed for this purpose, which comprises the states for both channels. As an example: if p=2 (binary data signal) and L=4 for both channels, the trellis diagram for the payload channel comprises 16 states, and the trellis diagram for the interference channel likewise comprises 16 states. The “combinational” overall trellis diagram which forms the basis for joint VA equalization of both signals then comprises 16×16=256 states. If one additional DF bit is taken into account in each case (that is to say L=5, LDF=4), the overall trellis diagram still comprises 256 states, but another two DF bits are added as the DF contribution to the calculation of the branch metrics.
The complexity for processing the overall trellis diagram is increased only by a factor of 16 in comparison to the complexity for processing the trellis diagram for the payload channel. When processing the trellis diagram by means of DSP (digital signal processor) control, a solution such as this leads to a very high MIPS load (MIPS: million instructions per second) on the DSP, so that other applications cannot run on the DSP, or can no longer run in an acceptable time. For a payload signal transmitted on the basis of the EDGE (Enhanced Data Rates for GSM Evolution) Standard (where p=8), equalization taking into account an interference source when using the overall trellis diagram is no longer possible in practical mobile radio use, owing to the excessively high DSP load.
In order to reduce the DSP load, it is already known for the processing of the trellis diagram to be assisted by specific dedicated hardware circuits, so-called hardware accelerators. These carry out the ACS operations in a trellis diagram time step by time step. In this case, the hardware accelerators can carry out very largely autonomous processing of the trellis diagram over a number of time units between two channel estimates (which result in a recalculation of the channel coefficients).
The German patent application with the file reference 103 23 407.4, which had not yet been published by the date of application, describes an equalization method in which a signal which is transmitted via a payload channel is equalized using the DF method and taking into account an interference channel. In this method, the trellis diagram for the payload channel and the trellis diagram for the interference channel are processed alternately in each time unit. The information about the interference channel (payload channel) obtained in the time unit under consideration is used as the DF contribution for the processing of the trellis diagram for the payload channel (interference channel) either in the same time unit or for the next time unit. One disadvantage of this procedure is that a specifically constructed hardware accelerator must be implemented owing to the change in the consideration of DF contributions from the respective adjacent channel.
The German laid-open specification DE 100 32 237 A1 describes a Viterbi equalizer which comprises a DSP as well as a hardware accelerator for carrying out the ACS operations. The hardware accelerator comprises a first hardware calculation circuit, which calculates partial sums of branch metric values, and calculates the branch metric values from new partial sums. A second hardware calculation circuit accesses the partial sums and branch metric values calculated in the first hardware calculation circuit, and carries out the ACS operations time step by time step. Reconfiguration of the hardware accelerator takes place only when a new channel estimate is produced, since the first hardware calculation circuit has to calculate new partial sums of the branch metric values, and the branch metric values from new partial sums, at this time.